High speed receivers circuits and methods

ABSTRACT

The present invention provides GPA embodiments. In some embodiments, a GPA stage with a negative capacitance unit is provided.

TECHNICAL FIELD

The present invention relates generally to high frequency receivers, andin particular, to gain-peaking amplifiers and equalization forhigh-frequency applications.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the invention are illustrated by way of example, and notby way of limitation, in the figures of the accompanying drawings inwhich like reference numerals refer to similar elements.

FIG. 1 shows a conventional composited gain peaking amplifier (GPA) withthree cascaded stages.

FIG. 2 shows a conventional gm-RL GPA stage for the composited GPA ofFIG. 1.

FIG. 3 is a diagram showing a single GPA gain stage in accordance withsome embodiments.

FIG. 4 shows a composited GPA amplifier formed from three cascaded GPAstages in accordance with some embodiments.

FIG. 5 shows a receiver with adaptive equalization and a compositedamplifier such as the amplifier of FIG. 4 in accordance with someembodiments.

FIG. 6 is a circuit showing a GPA stage of FIG. 3 in greater detail inaccordance with some embodiments.

FIG. 7 is a diagram showing an offset control topology for GPAs in acomposite amplifier in accordance with some embodiments.

FIGS. 8A-8C are diagrams showing offset-voltage detecting concepts forcontrolling an offset control topology in accordance with someembodiments.

FIG. 9 shows a truth table for GPA offset detection in accordance withsome embodiments.

FIGS. 10A and 10B illustrate first and second modes for shapingfrequency responses using a composite GPA in accordance with someembodiments.

FIG. 11 shows a circuit layout implementation for an LC-LC dualresonance circuit in accordance with some embodiments.

FIG. 12 shows an AC equivalent circuit for cascaded SDG-Gm and LC-Tiablocks in accordance with some embodiments.

FIG. 13 shows an AC-equivalent circuit for a negative capacitance unitand an input admittance in accordance with some embodiments.

FIG. 14 shows an AC equivalent circuit for cascaded SDG-Gm and LC-Tiablocks with the negative capacitance unit included therewith inaccordance with some embodiments.

FIG. 15 is a diagram showing a receiver with a composite GPA andedge-equalization with binning in accordance with some embodiments.

FIG. 16A is a diagram showing a zero-crossing histogram with a idealequalization in accordance with some embodiments.

FIG. 16B is a diagram showing a zero-crossing histogram with excessiveequalization in accordance with some embodiments.

FIG. 16C is a diagram showing a zero-crossing histogram withinsufficient equalization in accordance with some embodiments.

FIG. 17 is a table showing UI binning criteria in accordance with someembodiments.

FIG. 18 shows a table showing UI binning criteria in accordance withsome other embodiments.

DETAILED DESCRIPTION

Serial I/O interfaces are being driven at ever increasing rates. Forexample, chip;-to-chip channels may be operated at 28 Gb/s or evenhigher. Such channels have become more challenging for serial I/Odesigns because of the severe transmission-line loss and significantsignal reflections. It can be particularly challenging to design andimplement receiver amplifiers such as the gain peaking amplifiers (GPA)that are commonly used in high frequency serial I/O receivers. (A GPAmay also sometimes be referred to as an CTLE, continuous-time linearequalization amplifier.)

FIG. 1 shows a conventional composited gain peaking amplifier (GPA) withthree cascaded stages, and FIG. 2 shows a conventional GPA stage circuitimplementation. As indicated in FIG. 2, such prior GPA solutions may bedesigned with a Gm-RL topology. Unfortunately, such circuits haveseveral limitations. The available GPA Gain-Bandwidth product, anindication of the maximum speed capacity of an amplifier, is dominantlydetermined by the output RC time constant, i.e. RL*Cout, where Cout isthe output loading and total parasitic. The transconductance (Gm or gm)is proportional to the square-root of the term, IR*W/L (W and Lcorresponding to a utilized transistor's width and length,respectively). Thus considerable increments in the bias current, IR, anddevice size, W/L, may be required to make a substantial gm change.

Moreover, RL is also limited by the condition of the output DCcommon-mode level to ensure the sufficient saturation margin of thedifferential pair amplifier (the output DC=Vcc−RL*IR). Two cascadedidentical gain-stages give a bandwidth reduction of 36%, while threecascaded identical gain-stages give a bandwidth reduction of 48%.

For high frequency applications, designs have been modified (asindicated in FIG. 2) by replacing the RL with a series combination ofthe RL and an additional inductor. However, most of the afore mentioneddrawbacks are still applied to this derivative gm-RL topology.Accordingly, new approaches may be desired.

FIG. 3 shows a GPA stage in accordance with some embodiments. This GPAcircuit comprises a source-degenerative transconductance stage (SDG-Gm),a negative capacitance unit (Negative-Cap), and a trans-impedance stagewith LC resonant circuits (LC-Tia), coupled as shown. The negative cap.unit in each stage serves to cancel capacitance on the inside node atthe output of the SDG-Gm section, which allows for the gain of theamplifier stage to be boosted. This is in contrast, for example, withthe prior art GPA stage of FIG. 2, which uses an output voltage RL load.A GPA stage with an inter-disposed negative cap unit instead uses acontrolled device, e.g., NMOS device, as a current source with highoutput impedance.

In order to achieve a large (if not a maximum) gain peaking performance,a composited GPA may be formed from two or more of these stages cascadedtogether. For example, FIG. 4 shows three of these stages cascadedtogether in a Cherry-Hooper amplifier topology with a control signal(Vcnt) for controlling a gain parameter for improving the overallgain-bandwidth response of the entire amplifier. So, the composited(Cherry-Hooper type) amplifier of FIG. 4 is different from an amplifierformed from simply cascading together three of the prior art GPA stages.

FIG. 5 is a block diagram of a receiver with speed-enhanced equalizationtechniques employing GPA stages with negative capacitance units and withoffset and common-mode control, as disclosed herein. Functionally, adisclosed full speed gain-peaking amplifier (GPA) stage can provide thefirst stage of CTLE to better control opening of the data eyes and thussustain the adequate manipulations in the subsequent digitalequalization (e.g., DFE and CDR blocks). The GPA can be controlled tocompensate for the generally low-pass frequency response characteristicsof the incoming transmission channel and to mitigate againstinter-symbol-interference (ISI) effects by boosting up thehigh-frequency intensity of the input data, but also, by suppressing thelow frequency components where desired. Sufficient bandwidth andgain-peaking characteristics (i.e. gain magnitude and gain slopes versusfrequency) may be employed for good GPA designs.

FIG. 6 illustrates a possible circuit embodiment for a single GPAgain-stage of FIG. 3. Basically, the SDG-Gm and LC-Tia blocks are formedas an RC-degenerated amplifier with a Cherry-Hooper topology to supportthe high-frequency equalizations. The parallel negative-cap unit is usedto minimize the parasitic capacitance between the SDG-Gm portion and theLC-Tia block, and it further boosts up the AC performance of the GPAgain-stage. The LC-LC blocks function as feedback elements for invertersformed from Mp5/Mn3 and Mp6/Mn4. They correspond to resonant circuits inseries with one another (see, e.g., FIG. 11 for an exemplary IC chipimplementation).

In the SDG-Gm block, the variable capacitance (VarC) and variableresistance (VarR) are both used to control the receiver equalization. Asignal for controlling VarC determines the GPA AC gain-slope over theoperating frequency band. It is typically desirable to generate an ACresponse that matches the inversed transfer-function of thetransmission-line. The variable resistor (VarR) sets the lower-frequencygain and provides an adequate ratio of the maximum peak-gain to thelower-frequency gain. The probing terminal, vcm, between two resistorstrings of the variable resistor network (VarR) is employed for outputcommon-mode detection on the previous cascaded gain-stage.

As illustrated in the figure, the depicted negative-cap unit is formedfrom a cross-coupled NMOS circuit with a shunt capacitor. The neg. cap.unit serves to cancel out parasitic capacitance between the SDG-Gm andLC-Tia blocks. (See also FIGS. 12-14 for AC analysis of thenegative-cap. unit, alone, and integrated into the SDG-Gm and LC-Tiablocks.)

The NMOS devices (Mn1 and Mn2) are biased at a nominal DC current, buton the other hand are also controlled by the terminals Vos1 and Vos2 tocorrect the output offset voltage at the Vout of the LC-Tia output port.This offset correction scheme is done primarily (if not always) as soonas possible when the power supply is turned up and the receiver is in acalibration mode.

In the negative-cap block, the two P-type current mirrors (Mmr1 andMmr2) are used to bias the cross-coupled PMOS devices (Mp3 and Mp4) andare also used to adjust the DC level of the output common-mode voltage,Vout, at the LC-Tia output port. The Voctr signal controls the biascurrent of the negative-cap unit, and thus, controls the peaking gainand also the gain/bandwidth of the overall composite GPA amplifier.

In the LC-Tia block, a pair of CMOS inverters with local feedbacks(LC-LC across their inputs and outputs) is incorporated. The controlledresistor and a dual LC resonant circuit (e.g., the LC/LC unit of FIG.11) are exploited in the feedback path for thermal and process-variationcompensation and high-frequency gain peaking.

Different inductance and capacitance values may be chosen to obtain dualresonant frequencies at the LC/LC unit in order to broaden thegain-peaking characteristics for each of the GPA gain-stages. For thethree-stage GPA, three different resonant frequencies of the LC/LC unitsare designed with three different values of LC combinations so that theoverall AC gain-peaking characteristics can be optimized to match adesired inverse transfer-function of the transmission-line. (Anillustration of such gain-peaking, showing contributions by each of thethree cascaded gain-stages, can be seen in FIGS. 10A and 10B. FIG. 10Ashows a first mode used to shape the transfer response of the compositeGPA to inversely match the transmission line. FIG. 10B shows a secondmode, which simply maximizes a target frequency region. Note that thedashed line (for the targeted peak-gain frequency) is shifted slightlyto the left of the actual peak to account for PVT inconsistencies.)

In some embodiments, the LC-Tia block may be implemented with an LC/LCunit residing in the feedback path with a series resistor. To save thechip area, the two inductors in one individual LC/LC unit may beimplemented with a single differential inductor template (e.g., a layoutp-cell) as shown in FIG. 11. In this embodiment, each leg of theinductor is connected in parallel to the varactor C1 (or C2) as one-halfof the dual resonant LC/LC circuit.

FIG. 7 is a convenient representation of a composite GPA in accordancewith some embodiments. It shows how the output common-mode voltage ofeach gain-stage may be detected over the three subsequent gain-stages ofthe composite amplifier. In some embodiments, the same differential-paircircuits may be re-used as part of the common-mode feedback network inorder to avoid the extra loading on the high-speed data path. Shown inthis figure, a DC control approach for the output common-modestabilization may be used. This output common-mode feedback (CMFB)network may be designed based on (i) avoiding extra loading to thehigh-speed data path, and (ii) probing on a genuine circuit path withoutintroducing errors of device mismatch caused by the use of additionalCMFB circuitry. The offset voltage corrections can be done at the inputport of the first gain-stage, or can be corrected at each individualgain-stage.

FIGS. 8A through 8C present offset-voltage detecting concepts.Basically, the individual stage offset correction is performed primarily(if not only) in a power-on calibration cycle for the receiver, whereinthe output offset stage can be calibrated stage by stage. In a normaloperation mode of the receiver, the real-time offset-voltage of theentire GPA may be detected at samplers using the approach of the datatransition-edge (both rising/falling edges) distributions (presenting atthe eye diagram). An offset control routine may be operated in a digitalpart of the receiver (or elsewhere) based on FIGS. 8A-8C to controloffset (using Vos1, and Vos2 terminals in FIG. 6) to cause rising andfalling edge distributions to sufficiently align, as shown in FIG. 8A.(The alignment routine determines the offset polarity based on thedistribution analysis of the rising/falling edges versus the phaseinterpolated (PI) clock edges. The correcting signals (Vos1, and Vos2)may then be fed back to the input biasing circuitry for offsetcorrection. This offset voltage correction control is intended tooperate as a bang-bang scheme. The table of FIG. 9 is a truth-table forpossible conditions of data transition-edges versus offset voltagepolarities. With reference to FIGS. 6 and 7, the common-mode voltage ispicked up at Vocmm1 and Vocmm2 and fed into a low pass filter (LPF). Thecommon-mode control signal for a preceding stage is generated from theLPF output, from each subsequent stage, to control current levels in thenegative cap units. A statistically analyzed distribution indicates ifthe offset is positive, resulting in the differential Vos1-Vos2controlling the GPA stage to be more negative, and visa versa.

(Note that the one digital detection circuit (right-half section of FIG.5) can be used both for offset correction, as discussed up to now, aswell as for digital equalization, which is discussed later in thisdisclosure.)

With reference to FIG. 12, an AC analysis of the cascaded SDG-Gm andLC-Tia blocks, without including the negative-cap unit, will now bepresented. The transfer function and the effective bandwidth can bederived as shown herein, beginning with the first order AC transferfunction of the cascaded SDG-Gm and LC-Tia:

$\frac{Vout}{Vin} = {{{gm}_{SDG} \cdot {Zf}} - \frac{{gm}_{SDG}}{{gm}_{TIA}}}$Where${gm}_{SDG} = {{\frac{{gm}_{P}}{1 + {{gm}_{P} \cdot Z_{SDG}}}\mspace{14mu}{and}\mspace{14mu}{gm}_{TIA}} = {{gm}_{P\; 2} + {gm}_{N\; 2}}}$$Z_{f} = {R_{f} + ( {j\;\omega\; L_{1}{}\frac{1}{j\;\omega\; C_{1}}} ) + ( {j\;\omega\; L_{2}{}\frac{1}{j\;\omega\; C_{2}}} )}$$Z_{SDG} = {R_{SDG}{}\frac{1}{j\;\omega\; C_{SDG}}}$${{Noted}\text{:}\mspace{14mu} A{}B} = \frac{A \cdot B}{A + B}$

An approximated effective bandwidth (dominant-pole) without including anegative-cap unit may be expressed as:

$\omega \approx \frac{2{gm}_{TIA}}{C_{gs} + C_{out} + {{gm}_{TIA} \cdot Z_{f} \cdot C_{gd}}}$

Ω is less sensitive to Zf*Cout

FIG. 13 shows an AC equivalent circuit for the negative-cap unit. Theinput admittance, Yin is derived and can be expressed as an equivalentresistance, Req, and an equivalent capacitance, Ceq. The inputadmittance, Yin=Req+Ceq, and Req, and Ceq can be express as follows:

$R_{eq} = {\frac{- 2}{{gm}_{P}} \cdot \frac{1 + {( \frac{\omega}{\omega_{T}} )^{2}( {1 + \frac{C_{sdg}}{Cgs}} )^{2}}}{( \frac{\omega}{\omega_{T}} )^{2}( \frac{C_{sdg}}{Cgs} )( {2 + \frac{C_{sdg}}{Cgs}} )}}$$C_{eq} = {{\frac{C_{sdg}}{2} \cdot \frac{{- 1} + {( \frac{\omega}{\omega_{T}} )^{2}( {1 + \frac{C_{sdg}}{Cgs}} )}}{1 + {( \frac{\omega}{\omega_{T}} )^{2}( {1 + \frac{C_{sdg}}{Cgs}} )^{2}}}}\mspace{14mu}{and}}$$\omega_{T} = \frac{{gm}_{P}}{Cgs}$

FIG. 14 shows the two AC equivalent circuits combined. As can be seen,the AC performance for each individual GPA gain-stage can be improvedbecause of the incorporated negative-cap unit. The Ceq reduces theparasitic of the total Cgs (of both PMOS, and NMOS, and also the otheradditional parasitic). In addition, the Req presents as a negativeresistance which is also beneficial to decrease the input resistance ofthe LC-Tia, because the generated ac current signal from the SDG-Gm canbe more efficiently coupled into the LC-Tia block. Therefore, the GPAgain-stage can be further enhanced in its AC performance with theincorporation of the negative-cap unit.

In some embodiments, disclosed composite GPA circuits, designed inCherry-Hooper topologies, may have various benefits. For example, theycan support data-rate operations of at least 28 GB/s because theireffective bandwidths are less sensitive to output RC time constants.Therefore, such designs can be made as high bandwidth implementations,even though the Zf is designed as high impedance (or high resistance).In most cases, this will be an improvement as compared to conventionalGm-RL designs, in which the bandwidth is reversely proportional to theload (RL).

In addition, some designs may have a higher driving capability on acapacitive load. Some designs may also have less bandwidth reductionwhen their stages are cascaded. They also may have lower powerconsumption, e.g., because the designs may provide higher gain thanprevious designs, so there will be more margin for the trade-off betweenpower consumption and AC gain.

Also, in some embodiments, there may be less frequency range withsaturated gain. For example, the use of a double resonant LC/LC unitprovides a pointing gain response. Therefore, the frequency region forthe saturated-gain (small gain-slope region) may be substantially lessthan that of prior designs.

Also with some embodiments, there may be at least two availableoperational modes on the gain peaking adjustment. As shown in FIGS. 10Aand 10B, two gain-peaking controlled modes are available in the receiverequalization design, providing more flexibility in high-speed receiverdevelopment. Moreover, with some designs, digital offset voltagedetection, e.g. using statistical distributions of the data transitionedges, offset voltage may be detection can be made in a digital domain,providing a feasible approach that can provide improved immunitiesagainst PVT variations. These and other benefits may be provided fromvarious embodiments disclosed herein.

Digital Equalization Using Edge UI Binning

In the following sections and with respect to FIGS. 15-19,transition-edge binning techniques for digital equalization, e.g., for acomposite GPA as discussed above, will now be discussed. Techniquesdiscussed herein may be employed in receiver-based adaptivecontinuous-time linear equalizer (CTLE) amplifiers, such as the onesshown in FIGS. 5 and 15.

FIG. 15 is a top-level block diagram showing equalization (EQ)approaches in accordance with some embodiments disclosed herein. In afirst embodiment, edge-equalization in a CTLE configuration may requireonly one VGA gain control-loop, and in a second embodiment,edge-equalization in a CTLE structure may comprise two VGA gaincontrol-loops, one with peak-gain control and the other withlow-frequency gain control.

Since the first embodiment is also part of the second embodiment, itwill primarily be described. With reference to FIG. 15, the basicoperations of the edge equalization can be described in the followingmanner.

A transmitter (Tx) transmits the data signal through the channel(T-line) to the receiver or the VGA input. Due to the ISI effect, theeye diagram at the VGA input is degraded. In order to correctly processthe incoming data signal from the Tx, signal equalization is required toenhance the eye opening, by compensating the high frequency componentsof the signal. A VGA with source-degeneration topology is used toperform the waveform conditioning function. The data signal is thenenhanced on both of its amplitudes and transition-edge slopes, and inturn, the zero-crossing distributions of the pulse edges are alsoshifted.

With the depicted equalizer, a technique herein referred to as “binning”is employed. With binning, separate counters are used to count differentdata and edge samples that are characterized as 1 B (bit unit interval),XB, or otherwise. (A bit unit interval is the period for a single bit,i.e., the inverse of a detected, or presumed, bit rate. For example, ifa 2.5 GB/s scheme is assumed, 1 B would be 40 pico-sec. So, if an edgeis assessed as arriving 80 pico-seconds after the last edge, then itwould be classified as a 2B edge, a 160 pico-second edge would be a 4Bedge, and so on.) with the depicted digital detector, three up/downcounters are used: one for 1B edges, one for X (any integer) edges, andone for both 1B and XB edges.

FIGS. 16A through 16C illustrate counted edge distributions for idealequalization (16A), excessive equalization (16B) and insufficientequalization (16C). IN these distribution diagrams, falling edges,including edges from the 1-UI (1B) pulses, the falling edges from themulti-UI (x-UI or XB) pulses other than 1-UI pulses, and the overalledges (All-UI) from any pulses (i.e. combination of the 1-UI and x-UI)are included.

When the eye-diagram is ideal (FIG. 16A), no equalization is needed, andthe edge distributions of 1-UI, x-UI and All-UI will all be lined upwith each other. The center of the All-UI distribution should also bealigned to the PI-edge clocks. Basically, this will remain true from astatistical view point, even though the phase of the PI clock iscontinuously adjusted by the CDR. If the detection outcome is assignedas “−1” for the earlier edges compared to the PI-edge clocks, and as“+1” for the later edge cases, the detected edge distributions can bequantized by using the up and down counters (UDC). Ideally, the up/downcounters should give a 0-Count result for the All-UI edge distribution.

In FIGS. 16B and 16C, the edge distributions for different ISIconditions are illustrated. FIG. 16B shows the edge distribution in anover-equalized case where the UDC delta between All-UI and 1-UI, andalso All-UI and x-UI, are highlighted as the criteria for theidentification of an over-equalized condition for the CTLE loop. Thedetection criteria for identifying the under-equalized condition aresimilarly presented in FIG. 16C.

From the view points of the circuit operations, as shown in the blockdiagram of FIG. 15, two samplers are used (Data and Phase) for thesampling operations at the centers, and edges of the data pulses. Theresults of these samplers are then loaded into two registers for furtherprocessing.

The data and edge samples not only work as phase-detectors for CDR, butalso determine the corresponding edge-occurrence timing relationshipbetween data edges and the PI-edge clock. The table of FIG. 17 shows atruth table of this 1-UI versus X-UI binning criteria in accordance withsome embodiments.

In afore mentioned second embodiment (two control loops), two additionalamplitude-error samplers (Error-1 and Error-2 Samplers from FIG. 15) areincorporated. The corresponding binning criteria with theamplitude-error detection are shown in the table of FIG. 18.

In the preceding description and following claims, the following termsshould be construed as follows: The terms “coupled” and “connected,”along with their derivatives, may be used. It should be understood thatthese terms are not intended as synonyms for each other. Rather, inparticular embodiments, “connected” is used to indicate that two or moreelements are in direct physical or electrical contact with each other.“Coupled” is used to indicate that two or more elements co-operate orinteract with each other, but they may or may not be in direct physicalor electrical contact.

The term “PMOS transistor” refers to a P-type metal oxide semiconductorfield effect transistor. Likewise, “NMOS transistor” refers to an N-typemetal oxide semiconductor field effect transistor. It should beappreciated that whenever the terms: “MOS transistor”, “NMOStransistor”, or “PMOS transistor” are used, unless otherwise expresslyindicated or dictated by the nature of their use, they are being used inan exemplary manner. They encompass the different varieties of MOSdevices including devices with different VTs, material types, insulatorthicknesses, gate(s) configurations, to mention just a few. Moreover,unless specifically referred to as MOS or the like, the term transistorcan include other suitable transistor types, e.g., junction-field-effecttransistors, bipolar-junction transistors, metal semiconductor FETs, andvarious types of three dimensional transistors, MOS or otherwise, knowntoday or not yet developed.

The invention is not limited to the embodiments described, but can bepracticed with modification and alteration within the spirit and scopeof the appended claims. For example, it should be appreciated that thepresent invention is applicable for use with all types of semiconductorintegrated circuit (“IC”) chips. Examples of these IC chips include butare not limited to processors, controllers, chip set components,programmable logic arrays (PLA), memory chips, network chips, and thelike.

It should also be appreciated that in some of the drawings, signalconductor lines are represented with lines. Some may be thicker, toindicate more constituent signal paths, have a number label, to indicatea number of constituent signal paths, and/or have arrows at one or moreends, to indicate primary information flow direction. This, however,should not be construed in a limiting manner. Rather, such added detailmay be used in connection with one or more exemplary embodiments tofacilitate easier understanding of a circuit. Any represented signallines, whether or not having additional information, may actuallycomprise one or more signals that may travel in multiple directions andmay be implemented with any suitable type of signal scheme, e.g.,digital or analog lines implemented with differential pairs, opticalfiber lines, and/or single-ended lines.

It should be appreciated that example sizes/models/values/ranges mayhave been given, although the present invention is not limited to thesame. As manufacturing techniques (e.g., photolithography) mature overtime, it is expected that devices of smaller size could be manufactured.In addition, well known power/ground connections to IC chips and othercomponents may or may not be shown within the FIGS, for simplicity ofillustration and discussion, and so as not to obscure the invention.Further, arrangements may be shown in block diagram form in order toavoid obscuring the invention, and also in view of the fact thatspecifics with respect to implementation of such block diagramarrangements are highly dependent upon the platform within which thepresent invention is to be implemented, i.e., such specifics should bewell within purview of one skilled in the art. Where specific details(e.g., circuits) are set forth in order to describe example embodimentsof the invention, it should be apparent to one skilled in the art thatthe invention can be practiced without, or with variation of, thesespecific details. The description is thus to be regarded as illustrativeinstead of limiting.

What is claimed is:
 1. A chip, comprising: a gain peaking amplifier(GPA) stage including (I) a source-degenerative trans-conductance stage(SDG-Gm) having an input to be a GPA stage input; (ii) a trans-impedanceamplifier (Tia) having an output to be a GPA stage output; and (iii) anegative capacitance unit (NCU) to couple the SDG-Gm to the Tia.
 2. Thechip of claim 1, in which the Tia comprises at least one LC resonantcircuit.
 3. The chip of claim 2, in which the at least one LC resonancecircuit comprises a dual resonant (LC-LC) circuit coupled to the SDG-Gmas part of a negative feedback path from the Tia to the SDG-Gm.
 4. Thechip of claim 1, in which the negative cap. unit comprises (i)cross-coupled transistors each having an output, and (ii) a capacitorcoupled between the transistor outputs.
 5. The chip of claim 4, in whichthe Tia comprises first and second inverters, each having a dualresonant circuit (LC-LC) coupled between an input and output of theinverter.
 6. The chip of claim 5, in which the cross-coupled transistorsof the NCU has inputs that are coupled to the Tia inverter inputs. 7.The chip of claim 1, in which the GPA stage is one of three GPA stagescascaded together to form a composite GPA amplifier.
 8. The chip ofclaim 7, in which the NCU in each stage serves to cancel capacitance onan inside node at an output of the SDG-Gm circuit.
 9. The chip of claim8, in which the NCU in each stage as a controlled current source coupledas a load for an associated SDG-Gm.
 10. The chip of claim 7, in whichthe composite GPA is configured as a Cherry Hooper amplifier.
 11. Thechip of claim 7, in which the composite GPA has at least one control forcontrolling output offset for at least one of the GPA stages.
 12. Thechip of claim 11, in which the at least one control is to control acurrent level in the NCU for the GPA stage whose output offset that isto be corrected.
 13. The chip of claim 7, further comprising a digitaldetector to process rising edge alignment distributions to adjust outputoffset for one or more of the GPA stages.
 14. The chip of claim 7,further comprising a digital detector to process rising edge alignmentdistributions to adjust output offset for one or more of the GPA stages.15. The chip of claim 7, in which each stage has a Tia with anadjustable resonance circuit so that a frequency response for each stagemay be adjusted.